Electric power system

ABSTRACT

Disclosed is an electrical power system adapted to receiving power from a source of alternating current, typically a small alternator, and supplying a direct current potential to a load. This power system includes two stages, the first stage comprising switchable rectifying devices such as silicon controlled rectifiers which interact with the impedance of the alternator or other source of alternating current power as a switching regulator to create an intermediate direct current potential. 
     The second stage operates from this intermediate direct current potential to produce a slightly lower output voltage, and creates a more accurate regulation than is possible with the first stage alone, the second stage greatly reduces ripple and high frequency noise components contained on the output of the first regulator stage. The second stage serves as an active low pass filter tuned to a desired DC voltage. In combination these two stages produce light weight, highly efficient power systems while providing an output that is essentially pure DC so that load devices such as aircraft radios or electronic computers may be connected directly to the output with or without a battery in parallel with the output.

BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention relates to a rectifier regulator system adapted toreceiving power from a source of AC power such as a permanent magnet orrotating field alternator and supplying direct current to a load such asthe communications and navigation radios in an aircraft. This regulatormay take the form of a two stage device, the first stage being aswitching regulator, the second stage being a linear device in the formshown. These are adapted to work together, however each stage definesunique improvements over the prior art in their own field.

2. Description of Prior Art

Typical small aircraft power systems of recent manufacture contain anadaptation of an automotive-type alternator in which a rotating memberconsists of steel poles and a copper winding to produce the requiredrotating magnetic field. The power to this winding is supplied throughslip rings or similar devices. The stator which is stationary, usuallyis wound three phase and connected through diodes to the output terminalof the alternator which is connected to the aircraft battery. A solidstate regulator senses the charging voltage of the battery and controlsthe excitation current to the field either by changing the effectiveresistance value between the field and the battery or by switching theconnection between the battery and the field on and off at a high ratewith an on duty cycle depending on the field current required tomaintain the required output.

An alternator regulator setup of this type produces large amounts ofelectrical noise arising from arcing of the slip rings, switchingtransients in the field circuit and from recovery time transientsassociated with the diodes in the stator output circuit. It is generalpractice to use capacitive, or a combination inductive and capacitivefiltering, in the output of the alternator to somewhat reduce this noiselevel. Also the impedance of the battery in parallel with the outputbypasses much of this noise to ground. However, even with theseprecautions the remaining noise level is enough to be noticeable andtherefore degrades the performance level of modern avionics. Theresponse time of this type of regulator is long, compared to the periodof the AC output of the alternator winding. Therefore this type ofsystem cannot reasonably operate without a battery in the system.Removal of the battery, either intentionally to save weight in certainapplications or inadvertently because of failure of the battery orassociated wiring often results in overvoltage failures of avionics.

It should be noted it is common practice to turn off all radios in anaircraft before starting or stopping engines because of the transientswhich may be created by the alternators as well as other systems are ofsuch magnitude they may be destructive to avionics. Further, the brushesand slip rings associated with this system have limited life,particularly at high altitude. Mean/time before brush failure of somealternators used in general aviation aircraft today, when operated at20,000 feet, is below 50 hours. Further, the residual magnetism in thesealternators is often so low that if the engine is started, such as byhand propping, when the battery is completely discharged, the alternatorwill not start to charge the battery.

Permanent magnet alternators can be manufactured by currently knownprocesses to produce approximately the same output for approximately thesame weight and speed as the wound rotor machines; however, thepermanent magnet alternators have generally not found use in suchgeneral aviation airborne applications. One of the reasons is the lackof a suitable regulator. Alternators of this type are often employed onsuch vehicles as motorcycles, snowmobiles, and outboard motors. In thesecases, regulation is accomplished generally by use of switching devicessuch as silicon controlled rectifiers, either in series or in shunt, aspart of the network between the alternator and the battery. This type ofrectifier-regulator combination produces high values of electromagneticinterference commonly called radio noise, which is generally not aproblem in the application previously mentioned because these vehiclesdo not commonly have electronics susceptible to such noise as aircraftmust for communications and navigations.

The objectives and purposes of this invention are as follows:

(1) to produce a simple, light, inexpensive, and highly efficient systemto rectify and regulate the output of an alternator and to produce pureDC suitable for operating the electrical systems of an aircraft or othervehicle, so that this output may be compatible with the needs of solidstate electronics devices such as communication and navigation radios aswell as computer and guidance systems of the type found in airborneapplications;

(2) to eliminate the need for a battery completely or to permit thecontinuing operation of the electronics of a system with a battery incase of battery failure;

(3) to eliminate any transients upon the starting and stopping of thealternator, rectifier, regulator system which may damage the loadsconnected to the system;

(4) to produce a highly accurate output voltage unaffected by wideranges of load or speed of the alternator or by a wide range oftemperature;

(5) to produce a power system which will not be damaged by suddenchanges in load to include all load resistances from zero to infinityand that even with sudden steps in magnitude of the load will notproduce transients or overshoots in the output which might damage solidstate devices contained in the loads;

(6) to produce a power system in which the power semiconductor devicesare all attached electrically as well as thermally to the heatdissipating surfaces either of a grounded heat sink or one additionalisolated heat sink;

(7) to produce a power system capable of starting without a battery orany other source of external power, under even the adverse combinationof very low ambient temperature such as -65° C. and a load resistancecorresponding to several times lower than the resistance which wouldnormally be associated with the full power output of the alternator.This last stated purpose is particularly desirable since certain typesof airborne equipment such as the power supply sections of increasinglycommon radar systems present a negative impedance to the power systemthat supply their direct current input power; and because aircraft arefrequently required to operate at very low ambient temperatures.

SUMMARY OF THE INVENTION

This invention relates to an electrical rectification and regulationsystem comprising two stages and adapted to be connected to a source ofalternating current power, such as a permanent magnet alternator. Thefirst stage is a bridge circuit containing two diodes and two siliconcontrolled rectifiers. This first stage also contains the means forsensing the output of that stage for comparing it to a desired value andcontrolling the gate pulses to the silicon controlled rectifiers tomaintain voltage at the desired value. A capacitor across the output ofthe first stage reduces the magnitude of the AC or ripple components ofits output. The first stage is followed by a second stage which is anactive, low pass filter which may be set to a desired direct currentvoltage. The purpose of this stage is to provide more accurateregulation and/or lower AC ripple and noise output than is possible withthe first stage only.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows a circuit diagram of a preferred embodiment of thisinvention.

FIG. 2 shows the control or feedback loop for the first stage of thecircuit shown in FIG. 1 modified to work independently of the secondstage.

It shall be understand that even though FIGS. 1 and 2 are based on asingle phase full wave bridge, the teachings of this invention may beapplied to half wave as well and also to multiphase applications, suchas three phase power sources.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT (1) Electrical PowerSystem

A circuit for receiving alternating current from a source of power, suchas an alternator, and supplying direct current to a load is shown inFIG. 1. Values given in this description will be typical of a systemadapted to supply 40 amps or less to a 28 volt load; however, oneskilled in the art may adapt this invention easily to other voltage andcurrent levels. The values given are stated to aid in the understandingof the circuit and are not intended to restrict the scope of thisinvention.

The regulator receives power through terminals labeled T1 and T2 from ACpower shown as an alternator. Typically this may be a permanent magnetalternator consisting of a stationary stator surrounded by a steel rimto which are mounted a number of permanent magnets. However, many othersources of alternating power may be used, such as field excited machinesor a conventional power line such as a 120 volt, 60 cycle connected inseries with a suitable current limiting impedance such as an inductor.The following description will be based upon the use of a permanentmagnet alternator, the internal impedance of which is primarilyinductive. Some applications of this invention may require shieldingsuch as a grounded braid over the leads from the alternator to T1 and T2to eliminate the radiation of radio frequency energy from these leads.

Diodes D3 and D4 and silicon controlled rectifiers or other switchingdevices shown as SCR1 and SCR2 are connected in a bridge configurationbetween the alternator and a first stage filter capacitor labeled C5.Thus, the voltage across C5 may be controlled by the application ofappropriate gate signals to SCR1 and SCR2 up to the maximum voltage orcurrent output capability of the alternator.

Application to a 3 phase delta connected alternator could be made withfour additional components corresponding in connection and use to SCR1,R3, D1 and D3.

Since the alternator has a inductive internal impedance, conduction ofone of the SCR's (SCR1 or SCR2) produces a phase shift in the currentoutput of the alternator compared to what its open circuit voltage wouldbe. Therefore, if the other SCR is not gated on immediately on the endof the conduction of the first SCR, the forward voltage across thesecond SCR will rise quite rapidly, possibly exceeding the establishedmaximum rate of change of voltage with time for the silicon controlledrectifiers SCR1 and SCR2.

The magnitude of this rate of change of voltage with the time is reducedand controlled by capacitor C1 and resistor R1 shown connected acrossthe alternator. The value of C1 actually controls the rate of rise ofthis voltage, and the resistor R1 controls the transient from thedischarge of C1 when the next gate pulse occurs to SCR1 or SCR2. Thus,R1 serves to reduce the di/dt required of SCR1 and SCR2 and theresulting electrical noise or electromagnetic interference. Typicalvalues would be 30 ohms for R1 and 0.05 microfarads for C1.

In the typical embodiment described here, diodes D3 and D4 and SCR1 andSCR2 could all be 40 ampere, 200 volt devices. Resistor R3 and R19 areconnected from the gate to the cathode of SCR1 and SCR2 respectively;these resistors tend to stabilize the operation of the circuit at hightemperatures where significant leakage may exist in either SCR1 or SCR2.

It should be noted that in the embodiment of the invention shown, theanodes of SCR1 and SCR2 are electrically connected to ground for thepositive output system with negative output to ground. Since negativeground is the most common configuration in vehicular and aircraftapplications, this arrangement is highly desirable compared to groundingthe anodes of the diodes shown as D3 and D4. This is becausecommercially available diodes will operate at higher temperatures thancommercially available silicon controlled rectifiers, and because thecase or heat transfer surface of commercial SCR's is the anode; thus,the thermal impedance from the junction to the heat sink most generallygrounded is less critical on the diodes than the silicon controlledrectifiers.

Cathodes of diodes D3 and D4 and the positive terminal of capacitor C5are tied together to the point that will be henceforth referred to asthe output of the first stage and is labeled as point X.

Referring again to FIG. 1, Y refers to a circuit connection point whichwill be used in the following description of the gate control circuitryfor SCR1 and SCR2. Point Y is connected to the anode of a diode D1, withthe cathode of that diode connected to the gate of SCR1. Point Y is alsoconnected to the anode of the diode D2, with the cathode of diode D2being connected to the gate of SCR2. Therefore, if a current path iscreated from ground to point Y, current will flow through either D1 orD2 to which ever SCR gate, which at that instant, is more negative withrespect to its anode, that being the SCR which at that instant can beturned on to allow the flow of power from the alternator to point X.Once turned on either SCR1 or SCR2 remains on unitl its anode current isreduced to zero as a result of the alternator output waveform.

The flow of current from ground to point Y is allowed and controlled bya third SCR designated SCR3, and a resistor R2 connected in series withits anode to ground, the cathode of SCR3 being connected directly topoint Y. Thus when SCR3 is gated on either SCR1 or SCR2, whichever is atthat instant biased in the direction it can conduct will be gated on.For purposes of the following portion of this circuit description,assume that at a given instant the anode of SCR1 is positive withrespect to its cathode. Further assume that neither SCR is conducting.Therefore, SCR2 must be reversed biased since SCR1 and SCR2 cannot bothbe forward biased at the same time, thus at the instant in this example,if a gate pulse is supplied to SCR3 it will turn on. If the turn onspeed capability of SCR3 is much faster than that of SCR1 then thecurrent through SCR3, D1, and the gate of SCR1 will be limited by R2thus allowing relatively small and inexpensive components to be used forSCR3 and D1. However, it should be realized that with some componentparameters R2 may be omitted from the circuit. Typical value of SCR3 inthe example previously discussed would be a 200 volt, 1/2 amp device.Also typical would be; R2, 10 ohms, R3 and R19, 19 ohms.

As soon as SCR1 turns on voltage from its anode to cathode decreases toits forward voltage drop which typically would be between one and twovolts. This is larger than the sum of the forward voltage drop of SCR3,diode D1 and the gate voltage to fire of SCR1, thus forcing SCR3 to turnoff since its anode current is reduced to zero. If gate potentialsufficient to fire SCR3 is applied during the next half cycle, diode D2will similarly conduct, turning on SCR2, allowing power to flow throughSCR2 and D3 from the alternator to point X. This supplies a load currenteffectively flowing between point X and ground, said ground point alsobeing described by label T4.

Devices which may be typically used for SCR3 or have a gate current tofire of a few micro amperes and a gate voltage to fire of 0.5 to 0.8volts. Resistor R4 and capacitor C2 are connected between the gate andthe cathode junction of SCR3 to prevent this device being turned on fromlow level leakage or capacitive coupling respectively, inherent in thedevice which controls this circuit shown as transistor Q1. Q1 representsa control device capable of controlling relatively high voltages atrelatively low currents. It has three terminals, an input terminal, anoutput terminal, and a common input-output terminal. While it is shownas a PNP transistor, other devices may be substituted. The emitter isshown as the common input-output terminal and is connected to point X.The base is shown as the input terminal and the collector is the outputterminal. The collector is connected through a current limiting resistorR6, to the gate of SCR3. For the example being given here a transistorwith a peak collector current capability of 10 milliamperes or more anda collector base breakdown voltage of at least 200 volts would besuitable. However average power dissipation in this device would benegligible, under 10 milliwatts in this example. Representative valuesfor R6 and R4 would be both 10k ohms and for C2 would be 0.2microfarads. For purpose of description, the Base of transistor Q1 isdefined as point Z. A resistor R5 connects point Z to point Y, point Zis also connected to the negative output, shown as ground, through aresistor designated R10. Upon the initial startup of the power systemthe voltage of point X may be zero, however, it can be seen that as soonas the alternator produces voltage, the path through either D1 and R3,or D3 and R19 will produce an instantaneously negative potential atpoint Y compared to point X. Thus, current will flow through resistor R5and the base emitter junction of Q1, turning Q1 on. This occurs aspreviously mentioned with zero voltage on point X with respect to groundand before either SCR1 or SCR2 have been gated on. This will turn on Q1which will supply current through R6 to produce a gate signal to SCR3turning it on and as previously described, gate on SCR1 or SCR2 as isappropriate at that instant, thus creating a positive potential at pointX with respect to ground. As soon as this potential is created currentwill flow through resistor R10 further biasing transistor Q1 in the ondirection. For the example, a typical value of R5 would be 100K ohms andfor R10 would be 7K ohms. This arrangement has the further advantagethat a gate signal can be produced at the point where the anode voltageof either SCR1 or SCR2 just starts to become positive with respect toits cathode. This is because direct current voltage existing betweenpoint X and ground is available to supply the voltage drops in Q1, R6and the gate voltage to fire for SCR3, thus allowing the firing of SCR1or SCR2 very nearly at the crossover point from negative to positivevoltage on their anodes, respectively, compared to their cathodes. Thisminimizes the generation of unwanted high frequency transients which cangenerate radio frequency interference.

It should be apparent from the foregoing description that the controlleddevice Q1 is normally biased in the "on" direction by an effectivecurrent source supplied through resistors R5 and R10. R5 supplies thecurrent for startup and R10 predominates after startup.

It is the purpose of the second control device, labeled Q2, and shown asa PNP transistor, to remove this normal "on" bias from Q1 by creating atselected times, a low impedance shunt path from the emitter to the baseof Q1, or with respect to the points previously defined, from point X topoint Z. Q2 is shown with its emitter connected to point X, itscollector connected to point Z and its base or input terminal connectedto its emitter through a resistor shown as R7.

To further describe this circuit the control or base terminal of Q2 isdesignated as point W. The base emitter voltage necessary to turn on abipolar transistor is a known parameter and has a relatively predictablevalue that does not vary appreciably from one transistor to anotherwithin a given family, and has a quite predictable negative temperaturecoefficient. In most cases this temperature coefficient is about 2millivolts per degree celsius.

R7 would normally be chosen so that the current through it at this baseemitter saturation voltage is very large compared to the actual basecurrent required for the device to supply the collected currentrepresented by the current flow previously described through R5 and R10.Thus current flowing out of point W (such as through resistor R8) whenit reaches the magnitude that it will produce a drop across R7 equal tothe base-emitter saturation voltage of device Q2, will turn Q2 on, thiswill shunt the current going through R5 and R10 away from the base, orcontrol electrode, of Q1 turning that device off, in turn removing thegate signal from SCR3; this, in turn, removes the gate signal from SCR1and SCR2 to prevent more power from being delivered to point X until thecurrent out of the control point W through resistor R8 is reduced.

Point W is connected to the output terminal of the supply, shown as T3,through resistor R8, zener diode Z1 and zener diode Z2 in series. Thecommercial processes which produce relatively low voltage zener diodes,say for instance between 3 and 10 volts, produce components with normalcommercial tolerences of ±5 to ±20%. If these devices are operated inthe forward biased mode, the forward characteristic is typically that ofa silicon diode; however, the rate of change of voltage with a givenchange in current around a selected operating point is generally morepredictable and represents a lower dynamic impedance than that in mostnormal diodes. It also should be noted that this parameter, at a giventemperature is usually predictable with an accuracy of 2% even from onemanufacturing lot to another. Also a very predictable negativetemperature coefficient of this voltage exists which is useful in thiscircuit.

Because of the previously mentioned characteristics, Z1 and Z2, in theforward biased mode, are used as the major portion of a voltagereference. This reference is used to establish the amount by which theaverage voltage at point X exceeds the voltage on the positive outputterminal of this supply or T3. It will be understood that other voltagereference devices may be substituted.

This voltage difference between points X and T3 becomes the operatingvoltage across the second stage of this supply. The remaining portion ofthis voltage reference is the base emitter saturation voltage oftransistor Q2 when operating at a collective voltage of approximately0.6 volts and a collector current established by the Ohms lawrelationship between the voltage from X to ground and the value ofresistor R10. This voltage must be multiplied by, and can therefore besomewhat adjusted by (R7+R8)/R7. Values of the resistors used must besubstituted in the preceding equation, with the selection of R7sufficiently low such that the current throught R7 is very largecompared to the actual base current of Q2. Typical value of R7 might be60 ohms, for R8 30 ohms. Z1 and Z2 typically might be 1/4 watt zenerdiodes of a nominal 4 volt, 20% rating. These values would establish avoltage difference between the average voltage at point X and thevoltage at the output terminal T3 to approximately 21/2 volts at 25° C.with a negative temperature coefficient of approximately 6 millivoltsper degree centigrade. If the voltage at point X tended to rise abovethis 21/2 volt point conduction through R8, Z1, and Z2 would increase,turning on Q2. This in turn would remove the base drive from transistorQ1 which removes the gate drive from SCR3 and in turn from SCR1 andSCR2, preventing further power transfer from the alternator to point Xand the associated capacitor C5. Conversely, if the voltage at point Xtended to decrease below the example value of 21/2 volts positivecompared to terminal T3, the current through R8, Z1 and Z2 woulddecrease turning off transistor Q2. Therefore, transistor Q1 would beturned on, in turn supplying a gate signal to SCR3 and thus to SCR1 andSCR2 so that power could flow from the alternator to point X through therectifier bridge consisting of SCR1, SCR2, D3 and D4, preventing afurther decrease in the voltage at point X. The negative temperaturecoefficient of this voltage established between point X and the outputterminal T3 is highly desirable since it minimizes the creation of heatin the output stage at high temperature which is exactly the point wherefurther heat would tend to be destructive to semiconductor devices, butat the same time increases this voltage across this stage at very lowtemperatures where the saturation voltages of the transistor in theoutput stage tend to increase therefore requiring more voltage tomaintain proper operation.

The circuit composed of Q2, R7, R8, Z1 and Z2 can be expected tomaintain the voltage difference between point X and terminal T3 to aaccuracy of ± approximately 1/10 of a volt to account for variationsnormally expected in readily available inexpensive commercial componentsused in these locations without specific calibration or selection ofcomponents for each individual supply. This value is well withinreasonable tolerances for the function of this two stage regulator andtherefore this circuit satisfies a design objective of eliminating thenecessity for calibration of the inner stage voltage, that is thevoltage from point X to ground as an independent step from calibratingthe output voltage or voltage from terminal T3 to ground, said groundalso being labeled as terminal T4.

The effects of temperature on semiconductor device characteristics arewell known and described in the literature and will be apparent to oneskilled in applying such art as disclosed herein.

It should also be noted that the current through Z1 and Z2 used asreference devices is established by the base emitter saturation voltageof transistor Q2 and the resistor R7 at the regulation point, thusestablishing the voltage difference between point X and output terminalT3 independent of the voltage between the terminal T3 and ground orterminal T4. Thus, this voltage is not affected by an overload on theoutput terminals of the supply and does not tend to decrease with thisoutput voltage. The supply, is therefore capable of starting from aninitial condition with a very low load impedance connected, and thealternator initially standing still and accelerated from that conditionto an operating RPM, without any external voltage being supplied andwithout the necessity of specifying the voltage drop across Z1 and Z2 atlevels far below their normal operating point.

R8 has the additional function of controlling the transient currentthrough the base of transistor Q2 and Z1 and Z2 to within safe levels ifthe output terminals of the supply are suddenly shorted or overloadedwhile the supply is in operation.

Consistant with the example values being given, the filter capacitor C5is an electrolytic capacitor of 30,000 microfarads with a voltage ratingof 35 volts and the voltage from point X to ground could typically beexpected to regulate within ±1/4 of a volt from no load to the full loadthat the alternator will supply at a given speed, and from that minimumspeed that the alternator can supply the load to the maximum design rpm.For a 12 pole alternator and 40 amp maximum load, the peak to peak valueof the ripple voltage on C5 will be approximately 1 volt or lessdepending on speed and load. The function of the remaining components inthe FIG. 1 will now be described. They can be grouped together asforming the second stage of this two stage regulator. It is the purposeof this stage to further improve the regulation and/or decrease theripple and transient components appearing at the output terminal T3.Thus, the second stage functions as an active, low pass filter set to agiven DC voltage. The second stage can be most readily understood bydescribing it in two sections. The first section being composed oftransistors or amplifying devices Q5, Q6, Q7, Q8, Q9, Q10, Q11, and Q12and associated resistors R14, R15, and R17, diode D7, and capacitor C3.These devices taken together may be thought of as simulating a very highcurrent, very high gain, PNP transistor, with its emitter connected topoint X, its collector connected to terminal T3, and its base connectedto the point labled as V. It should be realized that bipolar transistorsare shown and used in the description of this circuit as the requiredamplifying devices; however, one skilled in the state of the art maysubstitute within the scope of this invention other types of amplifyingor control devices which in general terms would have an input terminal,an output terminal, and a common input-output terminal, generallycorresponding to the base, collector and emitter terminals respectivelyof the bipolar transistors shown.

The remaining components, R9, R11, R12, R13, R16, R18, D5, D6, Z3, Z4,Z5, Q3, and Q4, serve to create a stable voltage reference and tocompare the output voltage existing at terminal T3 with that voltagereference, and to either increase the control or base current to thepreviously mentioned transistor equivalent (point V) if the voltage atterminal T3 tends to decrease below the desired level, or to decreasethe drive current at that control or base terminal if the voltage atterminal T3 tends to increase above the desired level. Thus, a closed,frequency dependent feedback loop is created, resulting in an output ofaccurately regulated direct current with a minimum of ripple or noisecontent.

This voltage reference and comparison circuit, as shown, is constructedof discrete components. However, it will be realized by those skilled inthe state of the art that integrated circuits are available throughcommercial channels which can serve these functions. Integrated circuitshowever, are not as immune to transients which may be encountered undersome applications of this invention as the discrete component shown inFIG. 1 may be specified to be. It should also be realized that manyother arrangements of discrete components may be readily applied bythose skilled in the known state of the art to supply the requiredvoltage reference and comparison functions.

The voltage reference devices for this supply are shown as three seriesconnected zener diodes Z3, Z4, and Z5. Depending upon the spaceavailable, the degree of temperature compensation necessary, and thecost of the components and the regulation accuracy required these mightbe replaced by a single diode. Use of three each commercial type 1N825which is a 6.2 volt nominal, temperature compensated zener would beconsistant with the example values being given.

The current through these zener diodes Z3, Z4, and Z5 is normallycontrolled primarily by resistor R18 and also flows through diode D6.The voltage across R18 may be computed by subtracting the sum of thezener diode voltages and the forward voltage drop of diode D6, typically7/10 of a volt, from the design output voltage, in the example case 28volts. Application of Ohms law gives the required value of R18 for thedesign operating point of the zener diodes. In this example, 1.2K ohmswould be appropriate.

Since the current for the reference diodes is drawn through D6 from theoutput which is accurately regulated and contains low ripple component,current through and therefore voltage across the zener diodes isextremely stable. It is the purpose of resistor R16 to supply sufficientcurrent through and therefore sufficient voltage across the zener diodesto assure that the supply is capable of starting from an initialcondition with the alternator at rest and a low load impedanceconnected. The value of R16 might typically be 20 times the value ofR18. Diode D6 prevents the small current which flows through R16 underinitial starting conditions, from being diverted to output terminal T3.

The junction between R18 and cathode of zener diode Z3 is connected tothe input terminal of an amplifying device Q4 shown as the base of a NPNbipolar transistor. Transistor Q4 is connected in a configuration wellknown in the literature and to those skilled in this art, with anothersimilar amplifying device Q3, as a differential comparitor. The commoninput-output terminals of these amplifying devices, shown as emitters,are connected together and through a current limiting resistor R13 toground. Resistors R9, R11, and R12 are connected in series as a voltagedivider across the output, that is from terminal T3 to ground.

The values of R9 compared to the sum R11 and R12 are calculated to givea voltage at the junctions of R9 and R11 equal to the sum of thevoltages of the reference elements Z3, Z4, Z5, when the output voltageat terminal T3 is at its desired level. However, the readily availablecommercial tolerances in these components is not as tight as thetolerance which may be required for the calibration accuracy of thissupply. Therefore, resistor R12 would be selected to calibrate theindividual supply. The accuracy obtainable is quite high if theresistors are stable and the value of R12 is much lower than the valueof R11. Use of a relatively high reference voltage such as the 19 voltsin the example circuit minimizes the effects of any slight differencesin the characteristics of Q3 and Q4. Thus, in many applications of thisinvention a matched transistor pair will not be required for Q3 and Q4.The values consistent with the example being presented here would beR13-7K ohms, R9-4K ohms, R11-7.5K ohms, R12-1K ohms. Transistors Q3 andQ4 are operated at a collective voltage of approximately 10 volts, andin a collector current of 3 milliamperes or less.

The components surrounded by the dotted line on FIG. 1 comprise whatwill be refered to as the pass stage. This stage can be viewed as athree terminal device with an input terminal labeled point V and outputterminal connected to point T3 and a common input-output terminalconnected to point X. It is the purpose of this stage to supply thevoltage gain, current gain, and power dissipation capability requiredfor proper operation of the power supply.

An analogy may be made to a PNP transistor with point X connected to theemitter, point V connected to the base, and terminal T3 connected to thecollector. Proceeding with this analogy, it is the purpose of thisunique arrangement of components to have a very high current gaincombined with low emitter collector and emitter base saturationvoltages. This stage is shown in the figures as being composed ofbipolar transistors and associated resistors, a diode and a capacitor.However, it should be realized that other amplifying devices withsimilar functioning connections may be substituted and that if eachtransistor shown as a PNP were replaced by an NPN and each transistorshown as an NPN were replaced by an PNP the functioning of the circuitwould remain equivalent.

In the form shown, the circuit has three stages of gain. These are a lowcurrent stage, labeled Q5, which will be referred to as a predriver, andan intermediate current stage, labeled Q6, which will be referred to asa driver, with a high current output stage, shown as a parallelcombination of Q7 through Q12, which will be referred to as an outputstage. It should be realized that the output stage may contain only onedevice or a large number of devices in parallel depending on the currentrequirement. The following values will be appropriate for the 28 volt,40 amp regulator being used as an example to help in the understandingin this invention. Each transistor is shown with a resistor parallelwith the base emitter-junction, thus reducing the effect of transistorleakage currents, particularly at high temperature. Thus R15 isconnected across the base-emitter junction of the predriver stage, R14across the base emitter junction of the driver stage and R17 across thebase emitter junction of the output stage. Example values would be 1000ohms for R15, 150 ohms for R14, and 15 ohms for R17.

A capacitor C3 is also shown connected between the base and emitter, orinput and common input-output terminals, of the predriver stage. It isthe purpose of this capacitor to give a reduction, or rolloff, of gainwith frequency so as to prevent instabilities or unwanted transientovershoots at high frequency. Commercial transistor types 2N6101 wouldbe appropriate for Q7 through Q12, 2N6491 for Q6 and 2N6111 for Q5.These devices allow sufficient surge and steady state current capacityto prevent damage even if the power supply was suddenly and repeatedlyshort circuited. In the short circuit condition the steady state currentwould be limited by the AC source shown as an alternator in FIG. 1.

In the following description the high current stage will be referred toas a single transistor Q7 even though as previously described it may bea parallel combination of devices. Point X is connected to the collectorof NPN transistor Q7 and also to the emitter of PNP transistor Q6. Thecollector of transistor Q6 is connected to the base of transistor Q7 thebase of Q6 is connected to the emitter of PNP transistor Q5, thecollector of transistor Q5 is connected to the emitter of transistor Q7and to the terminal T3. The base of transistor Q5 is connected to pointV.

One knowledgable in the ratings of devices of this type, which are quitetypical of a large number of commercial devices in use today, willrealize that the emitter-base saturation voltage of transistor Q6 plusthe emitter-collector saturation voltage of transistor Q5 (because ofthe relatively low current through these paths) will be lower than thesum of the emitter-collector saturation voltage of Q6 and theemitter-base saturation voltage of Q7. Since these two paths areconnected in parallel, the effective saturation voltage between point Xand terminal T3 will be emitter-collector saturation voltage of Q6 plusthe emitter base saturation of transistor Q7. 1.0 volts at 40 amps istypical for the device types given above. Therefore, Q5 will beoperating at a collector voltage above the saturation value and thus inits active region. In the foregoing discussion, magnitude should betaken numerically as absolute values. The input saturation voltage ofthe circuit, or the voltage from terminal X to terminal V, can be seento be the sum of the base-emitter saturation voltage of transistor Q6plus the base-emitter saturation voltage of transistor Q5.

The current gain of the circuit will be the product of the current gainof the three stages, that is, h_(FE) of Q5 times h_(FE) of Q6 timesh_(FE) of Q7, with gain measured at the appropriately operating currentfor each stage. Allowance would also have to be made for the currentflowing through the stabilizing resistors R14, R15, and R17.

Comparison will now be made with the Darlington transistorconfiguration. The Darlington configuration will not be shown ordescribed in detail here since it is well described in the literatureand understood by those working in this field. It is assumed that theDarlington is made from either a pair of NPN cor a pair of PNPtransistors. This is the usual method. The Darlington transistorconfiguration has a base-emitter saturation which is the sum of thebase-emitter saturations of two transistors and a collector-emittersaturation equal to the sum of the base-emitter saturation of onetransistor and the collector-emitter saturation of another transistor.This is exactly the same as just described for the circuit shown withinthe dotted line, the difference being that the circuit shown as part ofthis invention has the gain of one additional transistor stage withoutthe increase saturation voltages normally associated with adding of suchan additional stage.

Thus, the circuit shown within the dotted line may be utilized toadvantage in many applications requiring a high gain, low saturationvoltage amplifying stage. This is particularly advantageous in a powersystem of this type since the voltage at point X needs to be set onlyhigher than the voltage required at the output terminal T3 by thesaturation voltage of the amplifying device connected between theseterminals plus a small additional voltage to assure that the device isoperated in its active region.

It should be noted as an object of this invention that not only theoutput but also the driver and predriver stages are operated at or nearsaturation in the conditions of severe overload or shortcircuit of theoutput of the supply, thus minimizing the heating and the chance offailure in these devices. To further minimize the dissipation in thedevices upon short circuit of the output, it should be noted that withthe short circuit on the output, resistor 9 is effectively in parallelwith the series combination of resistors 11 and 12, thus presenting arelatively low impedance from the cathode of diode D5 to ground. Thiscombination then effectively parallels resistor R13, increasing theemitter current that may flow through transistor Q4 to supply therequired drive to the predriver stage Q5. Thus, protection from overloadis supplied without resorting to low values of R13 which would result inhighly imbalanced currents in Q3 and Q4 under operating conditions intoa normal load. D5 further protects the base-emitter junction oftransistor Q3 from high and possibly destructive inverse voltages whichmight otherwise exist as a transient upon sudden application of a shortcircuit from terminal T3 to T4. Typical value for capacitor C3 for theexample being given would be 0.5 microfarrods. If the intendedapplication of the supply may include connection of the output to abattery or connection of the output to a source auxilary or groundpower, a diode shown as D7 may be desirable. If the connection to theexternal source of power is made before the alternator is producingoutput, capacitor C5 will not be charged; thus, the voltage connected tothe output would appear in the reverse direction across the base-emitterjunction of transistor Q7, and could be of such magnitude as to damagethe transistor. The diode, shown as D7, across this junction bypassesthis current to the base of transistor Q7. The current may then flowfrom the base to the collector terminal since that junction is forwardbiased. It should be noted as an object of this diode placement thatthis current flowing through the base-collector junction will beamplified in the transistor, resulting in an additional current flowthrough the reverse biased emitter-base junction. Thus, in effect, thecollector is acting not as a collector but as an emitter, and theemitter is acting not as an emitter but as a collector. It is known thatthe gain of a transistor in this reverse-connected mode is relativelylow; however, it is sufficient to considerably reduce the size requiredfor diode D7 compared to an alternate location which would for its anodeas shown to terminal T3 but its cathode to an alternate locationdirectly to point X.

Capacitor C4 is connected directly across the output. It serves thefunction of an output filter capacitor used in most power supplies. Inthe example being given its value might range from a fraction of amicrofarad to several thousand microfarads, depending on the nature ofthe load and temperature range over which the power system was designedto operate.

To complete the example being given for the design of a 40 amp, 28 voltoutput between terminal T3 and T4, 28 volt may be expected to varyapproximately 10 millivolts from no load to full load and over the rangeof alternator speeds giving sufficient output to carry the load. Outputripple can be expected to be in the order of one millivolt or less.Voltage at point X could be expected to be approximately 30 volts with aripple of approximately one volt peak to peak and a regulation at thatpoint of approximately + or -2/10 of a volt over a range of load andspeed.

The efficiency of the supply would be approximately 90% at full load andefficiency would not decrease appreciably with the output currentdecreased. This high efficiency at low output current compared to therated load of the supply, is a highly desirable and unique advantage ofthis invention. It arises from the very low power dissipation necessaryin the control circuits associated with transistors Q1, Q2, Q3 and Q4,and their associated components. Thus, almost the entire powerdissipation is in the power bridge circuit SCR1, SCR2, D3 and D4 and inthe transistor stage Q7, thus as the output current decreases, the powerloss in these components decrease. This is an advantage over the usualswitching power supply which may exhibit high efficiency at high loadbut generally is known to have relatively low efficiency at decreasedload.

Operating from a given alternator, a supply of the type described hereinhas been shown to be capable of delivering a given load power at a speedlower than that obtainable by a high frenquency switching regulator.This comparison of the minimum alternator speed and therefore outputthat will supply a given DC load is the most critical measure of supplyefficiency in many systems.

FIG. 2 in most respects shows a circuit which operates in the same wayas the circuit shown in FIG. 1. All components with similar numbersserve the same purposes and thus will not be described again. Fig. 2differs in that the reference for the interstage voltage, that is thevoltage between point X and ground, is established by a reference toground as opposed to a reference to the output terminal T3 as is shownin FIG. 1. Thus, Z1 and Z2, which in the forward biased mode of FIG. 1,completed the control loop of the first stage to the output of thesecond stage are replaced by a single zener labeled Z6. Z6 is connectedfrom the end of resistor R8 to ground. Z6 is operated in the zenerregion and thus would be selected to have a zener breakdown voltage ofapproximately one volt below the desired voltage at point X. If it isdesired to establish voltage at point X more accurately than iseconomical with the component placement shown in FIG. 2, resistor R8could be eliminated, the cathode of zener Z6 would then be connecteddirectly to the base of transistor Q2, the anode of Z6 would then beconnected to the center point of a series combination of two resistors,with the ends of that resistor voltage divider connected to point X andground. Thus, the voltage at point X could be accurately controlled byselecting or adjusting one of these last two described resistors. Therequired voltage for Z6 would thus be reduced by approximately the ratioof the voltage divider. It would also be possible since the temperaturecoefficient of zener diodes is known to be predictable as a function ofthe zener voltage, to combine this with the known temperaturecoefficient of the base-emitter saturation voltage of transistor Q2, toproduce a known temperature coefficient of the voltage appearing frompoint X to ground. The circuit, as shown either in FIG. 2 or as justdescribed, allows the operation of the first stage completelyindependent of the operation of the second stage, and therefore in somenoncritical applications the second stage of the supply could beeliminated completely. It should be realized that the functionsattributed to Z6 and its associated resistors could be replaced by anycombination of devices which effectively produce an increasing basecurrent in transistor Q2 when the voltage from point X to groundincreases above a preselected value.

I claim:
 1. An electrical system, said system comprising:a source ofalternating current power, the frequency, voltage and waveform of whichfluctuate with respect to time and having an impedance to limit themaximum current available therefrom; first circuit means coupled to saidsource and containing at least one device normally nonconductive in agiven direction, and which may be rendered conducting in said directionresponsive to a control signal applied thereto; means for supplying saidcontrol signal to said device until the voltage output of said firstcircuit means reaches a desired level, and removing said control signalwhen said first circuit means output rises above said level; and anactive, low pass filter with a feedback loop set to a desired directcurrent voltage, and with an input terminal coupled to the output ofsaid first circuit means, the output of said active, low pass filteradapted to supply direct current to said load.
 2. The system recited inclaim 1 wherein said device comprises a controlled rectifier.
 3. Anelectrical power system comprising:at least two three-terminal solidstate devices of the type normally not conducting in a given directionbut that may be rendered conducting between first and second terminalsin response to a control signal applied between the third terminal andthe second terminal, with the first terminals of both said deviceselectrically connected, and the second terminals of said devices coupledto a source of power flow, the third terminal of each device connectedthrough rectification means to a common control point; an additionalthree-terminal solid state devices normally not conducting in a givendirection and which may be rendered conducting in said direction betweenfirst and second terminals thereof in response to a control signalapplied between the third terminal and second terminal, the firstterminal of said additional devices coupled to said common connection ofsaid first terminals of said first two devices, and the second terminalof said additional device being coupled to said common control point. 4.An electrical circuit comprising:first, second and third three-terminalcircuit devices, each of said first, second and third devices includingan input terminal, an output terminal, and a common input-outputterminal; said second circuit device having its common input-outputterminal coupled to the output terminal of said first device, and itsoutput terminal coupled to the input terminal of said first device; saidthird circuit device having its common input-output terminal coupled tothe input terminal of said second device, and its output terminalcoupled to the common input-output terminal of said first device; andcapacitive means effectively coupled between said common input-outputterminal and the input terminal of said third circuit device.
 5. Theelectrical system recited in claim 1 wherein said source of alternatingcurrent comprises and engine-driven alternator, the speed of whichvaries with engine speed.
 6. The electrical system recited in claim 5wherein said engine-driven alternator comprises a permanent magnetalternator.
 7. The electrical system recited in claim 1 wherein saidfirst circuit means comprises two or more control rectifiers, eachhaving cathode, anode and gate terminals.
 8. The electrical systemrecited in claim 7 further comprising a diode connected between eachgate of each control rectifier and the means for supplying said controlsignal.
 9. The electrical system recited in claim 7 further comprisingmeans for producing substantial voltage between the cathodes of saidcontrol rectifiers.
 10. The electrical power system recited in claim 3or 5 wherein said three-terminal devices comprises silicon controlrectifiers.
 11. The electrical power system recited in claim 4 whereinthe output terminal of said third device is directly connected to thecommon input-output terminal of the first device.
 12. The electricalpower system recited in claim 4 wherein the first circuit device carriesthe highest current level, the second circuit device carries theintermediate current level, and the third circuit device carries thelowest current level.
 13. An electrical circuit comprising:first, secondand third three-terminal circuit devices, each of said first, second andthird devices including an input terminal, an output terminal, and acommon input-output terminal; said second circuit device having itscommon input-output terminal in direct electrical connection to theoutput terminal of said first device, and its output terminal coupled tothe input terminal of said first device; said third circuit devicehaving its common input-output terminal coupled to the input terminal ofsaid second device, and its output terminal coupled to the commoninput-output terminal of said first device; and wherein said directelectrical connection of the common input-output terminal of said secondcircuit minimizes the voltage drop between the output and commoninput-output terminals of said first device, and which produces acircuit having the current gain of all three of said circuit deviceswhile maintaining a very low saturation voltage between thecurrent-carrying output and common input-output terminals of said firstdevice.
 14. A system for controlling power transfer from an alternatorto a direct current load comprising:(a) an alternator; (b) a directcurrent load; (c) at least one semiconductor having a control electrodedevice and capable of blocking the flow of current in a given directionbetween said alternator and said direct current load until theapplication of a signal at said control electrode and thereafterallowing the flow of current in said direction; (d) a first amplifyingdevice with an input terminal, an output terminal and a commoninput-output terminal, said output terminal coupled to control thesignal to said control electrode, and said input terminal connected tobe biased on by a signal relative to said common input-output terminalby a portion of the output from said alternator; and (e) another circuitdevice connected to remove said on-biased signal in response to aselected condition of said direct current load.
 15. The system recitedin claim 14 wherein said another device is an amplifying deviceconnected to said direct current load through voltage reference means.16. The system recited in claim 14 further comprising:(a) a currentcontrol network; and wherein (b) said input terminal is connected tosaid current control network for providing a selected level for theon-biased signal from said portion of the alternator output.
 17. Thesystem recited in claim 16 wherein said current control network is aresistor.
 18. The system recited in claim 16 wherein said currentcontrol network is a resistor and a rectifier in series with saidresistor.
 19. A system for supplying power from a permanent magneticalternator to a direct current load comprising a silicon-controlledrectifier (SCR), a first transistor having an input terminal, and acurrent control network, with said SCR connected to control the flow ofcurrent from said alternator to said load, the gate terminal of said SCRbeing connected to respond to the output of said first transistor, theinput terminal of said first transistor connected through said currentcontrol network to render said first transistor conducting during aselected portion of said alternator cycle, and bypass circuit meanscoupled with said system to bypass the output of said current controlnetwork in response to a preselected voltage level of the direct currentload.
 20. The system recited in claim 19 wherein said bypass circuitmeans comprises a second transistor and a voltage reference element,said second transistor coupled to bypass the output of said currentcontrol network in response to an input signal to said second transistorderived from the voltage across said direct current load minus thevoltage across said voltage reference element.